1. Field of the Invention
The present invention relates to a quadrature amplitude modulator with distortion compensation to be used in a communication such as a radio communication.
2. Description of the Background Art
In a communication such as a radio communication, a modulator is used for generating narrow band signals such as multi-level PSK (Phase Shift Keying) and QAM (Quadrature Amplitude Modulation).
In particular, in a case of digital modulation and demodulation schemes to be utilized in digital mobile communications, a modulator with higher accuracy is required compared with analog modulation and demodulation schemes because the digital schemes utilize waveform transmission instead of spectral transmission of the analogue schemes. For this reason, such a digital modulation and demodulation scheme uses a quadrature modulator to which an in-phase amplitude and a quadrature amplitude are applied as baseband signal inputs. However, a conventional quadrature modulator has the following problems.
(i) A carrier leak component is superposed onto the modulated wave even when the off-set levels of the baseband signal inputs are balanced, because it is difficult to balance the baseband signal inputs as the effective inputs inside the quadrature modulator.
(ii) An image component is superposed onto the modulated wave even when the amplitudes of the baseband signal inputs are balanced, because it is difficult to balance the baseband signal amplitudes as the effective inputs inside the quadrature modulator.
(iii) The quadrature modulator requires two input carrier signals with 0 phase and .pi./2 phase. It is, however, difficult to manufacture an accurate phase shifter for generating the .pi./2 carrier signal on an IC. When this orthogonal phase relationship between two carrier signals is not accurate, an image component is superposed onto the modulated wave.
(iv) When the quadrature modulator is implemented on an IC, its operational characteristics vary according to the variations of the power source voltage and the surrounding temperature.
In conventional quadrature modulators, these problems are solved by skillful adjustments of manufacturers. As for the variation due to aging, the necessary adjustment is provided by a regular maintenance service.
In order to adjust such a distortion compensation of the quadrature modulator automatically, two simple configurations have been proposed conventionally.
The first proposition is a configuration for compensating the distortion due to the inaccurate orthogonal phase relationships between the baseband signal inputs, disclosed in Japanese Patent Application Laid Open No. 62-13143, which is schematically shown in FIG. 1.
This quadrature modulator of FIG. 1 comprises: an in-phase amplitude input terminal 40 for receiving an in-phase amplitude input signal I(t) in baseband; a quadrature amplitude input terminal 41 for receiving a quadrature amplitude input signal Q(t) in baseband; a modulated wave output terminal 42 for outputting a modulated wave obtained from the baseband signal inputs; an oscillator 46; a .pi./2 phase shifter 45 for shifting the phase of the carrier generated by the oscillator 46; a first mixer 43 for mixing the in-phase amplitude input signal I(t) entered from the in-phase amplitude input terminal 40 with the phase shifted carrier generated by the .pi./2 phase shifter 45; a second mixer 44 for mixing the quadrature amplitude input signal Q(t) entered from the quadrature amplitude input terminal 41 with the carrier generated by the oscillator 46; a combiner 47 for combining the outputs of the first and second mixers 43 and 44; a signal identifier 48 for identifying signal constellation points constituted by the combinations of the in-phase amplitude input signal I(t) and the quadrature amplitude input signal Q(t); and an amplitude calculator 49 for measuring amplitudes of the signal constellation points identified by the signal identifier 48, obtaining a difference or a ratio of the measured amplitudes, and adjusting the .pi./2 phase shifter 45.
The measured amplitudes do not coincide with each other unless the .pi./2 phase shifter 45 is operating accurately, so that the compensation of the distortion due to the inaccurate orthogonal phase relationships between the carriers can be made automatically by adjusting the operation of the .pi./2 phase shifter 45 according to the amplitude difference obtained by the amplitude calculator 49.
However, this quadrature modulator configuration of FIG. 1 is associated with the following drawbacks.
(i) A fine adjustment of the phase shifter required in this configuration is difficult to realize in a practical phase shifter circuit.
(ii) A practical procedure for controlling the phase shifter is not disclosed.
(iii) An error due to the roll off shaping occurs in practice so that the practically achievable accuracy is insufficient.
(iv) The achievable accuracy becomes insufficient under the presence of the other distortions such as those due to a DC off-set and an amplitude imbalance.
The second proposition is a configuration for compensating the distortion due to the DC off-set, disclosed in Japanese Patent Application Laid Open No. 63-62439, which is schematically shown in FIG. 2. In FIG. 2, those elements which are equivalent to the corresponding elements appeared in FIG. 1 above are labelled by the same reference numerals.
In this quadrature modulator configuration of FIG. 2, the in-phase amplitude input signal I(t) applied to the in-phase amplitude input terminal 40 is mixed with a carrier generated by the oscillator 46 at the first mixer 43 and then the DC off-set in the output of the first mixer 43 is calculated by a DC off-set calculation unit 50, in which the output of the first mixer 43 is rectified and integrated for a negative modulation symbol and a positive modulation symbol separately according to a modulation symbol detected by a comparator 51 which compares the entered in-phase amplitude input signal I(t) with a ground level to determine the modulation symbol. The integrated values for the negative and positive modulation symbols are then added together to yield the total DC off-set which is negatively fed back to the in-phase amplitude input signal I(t) inputted into the first mixer 43 by a feed back loop 52 such that the DC off-set can be compensated. The DC off-set for the quadrature amplitude input signal Q(t) is compensated in the similar manner.
However, this quadrature modulator configuration of FIG. 2 is associated with the following drawbacks.
Namely, in this configuration of FIG. 2, the compensation of the distortion due to the DC off-set is carried out for the in-phase amplitude input signal I(t) and the quadrature amplitude input signal Q(t) separately, but this compensation of the distortion due to the DC off-set before the combiner 47 is actually insufficient because the further distortion is caused by the superposition of the local carrier which leaks in an actual high frequency circuit.
In addition, this configuration of FIG. 2 requires a considerably complicated circuit configuration.
In general, the modulated wave obtained by a quadrature modulator is impaired by a linear distortion as well as a nonlinear distortion. The nonlinear distortion is caused by a nonlinear response of diodes used in the mixers in the quadrature modulator, which can be suppressed relatively easily by reducing the input levels appropriately. On the other hand, the compensation of the linear distortion requires selection of well balanced parts or fine adjustment. This situation concerning the linear distortion will now be described in detail by using the quadrature modulator configuration of FIG. 1 described above as an example.
Namely, at the first and second mixers 43 and 44, the in-phase amplitude input signal I(t) and the quadrature amplitude input signal Q(t) are multiplied by a carrier wave r.sub.C (t)=cos (.omega..sub.C t) of an angular frequency .omega..sub.C generated by the oscillator 46 and a phase shifted carrier wave r.sub.S =-sin(.omega..sub.C t) obtained by the .pi./2 phase shifter 45, respectively, and then combined together at the combiner 47. However, a practical quadrature modulator is affected by the carrier leak due to the stray capacitance or the stray inductance as well as by the image generation due to the phase shift error by the .pi./2 phase shifter 45.
Consequently the actual modulator output y(t) is expressed by: EQU y(t)=y.sub.1 (t)+y.sub.2 (t)+y.sub.3 (t)+y.sub.4 (t) (1)
where y.sub.1 (t) is a response for a case the DC off-set .delta..sub.C1 is added to the baseband signal at the first mixer 43, which is given by: EQU y.sub.1 (t)=[I(t)+.delta..sub.C1 ] cos (.omega..sub.C t) (2)
y.sub.2 (t) is a response for a case in which the baseband signal with the DC off-set .delta..sub.S1 added is multiplied with the quadrature carrier of a phase shift error .theta. as the gain of the second mixer 44 is .alpha. times that of the first mixer 43, which is given by: EQU y.sub.2 (t)=[-.alpha.Q(t)+.delta..sub.S1 ] sin (.omega..sub.C t+.theta.)(3)
y.sub.3 (t) is a response for a case in which an amplitude and a phase of the carrier leak for the in-phase component are .delta..sub.C2 and .theta..sub.1, respectively, which is given by: EQU y.sub.3 (t)=.delta..sub.C2 cos (.omega..sub.C t+.theta..sub.1)(4)
and y.sub.4 (t) is a response for a case in which an amplitude and a phase of the carrier leak for the in-phase component are .delta..sub.S2 and .theta..sub.2, respectively, which is given by: EQU y.sub.4 (t)=.delta..sub.S2 sin (.omega..sub.C t+.theta..sub.2)(5)
Thus, the modulator output y(t) can also be expressed as: EQU y(t)=c(t) cos (.omega..sub.C t)-d(t) sin (.omega..sub.C t) (6)
where c(t) is an in-phase modulated signal which is given by: EQU c(t)=I(t)+.delta..sub.C +sin .theta.[-.alpha.Q(t)+.delta..sub.S1 ](7)
and d(t) is a quadrature modulated signal which is given by: EQU d(t)=-.alpha.Q(t) cos .theta.+cos .theta..multidot..delta..sub.S1 +.delta..sub.S ( 8)
in which the off-sets .delta..sub.C and .delta..sub.S are given by: EQU .delta..sub.C =.delta..sub.C2 cos .theta..sub.2 +.delta..sub.S2 sin .theta..sub.S2 sin .theta..sub.2 ( 9) EQU .delta..sub.S =-.delta..sub.C2 sin .theta..sub.1 +.delta..sub.S2 cos .theta..sub.2 ( 10)
However, the actual output of a conventional quadrature modulator such as those described above is not an ideal one in which the in-phase amplitude input signal I(t) and the in-phase modulated signal c(t) are equal and the quadrature amplitude input signal Q(t) and the quadrature modulated signal d(t) are equal.
Namely, as shown in a signal space diagram of FIG. 3, in an ideal case, the Lissajous' figure for the output of the quadrature modulator with the in-phase amplitude input signal I(t)=cos (x) and the quadrature amplitude input signal Q(t)=sin (x) for x=0 to 2.pi. appears as a circle indicated by a solid line in FIG. 3, but the actual Lissajous' figure for the output of the actual quadrature modulator appears as an oblique oval with a center shifted away from an origin which is indicated by a broken line in FIG. 3 because of the linear distortions. In this Lissajous' figure for the actual output, the normal figure shown in FIG. 4A is superposed with the figure involving the DC off-set shown in FIG. 4B, the figure involving the amplitude imbalance shown in FIG. 4C, and the figure involving the inaccurate orthogonal phase relationship shown in FIG. 4D, where the deviation from the origin represents the carrier leak and the deviation from the circular shape represents the image. Thus, in the Lissajous' figure for the actual output shown by the broken line in FIG. 3, the contributions from the distortions shown in FIGS. 4B, 4C, and 4D are mixed up.
In order to compensate such a linear distortion, it is necessary to carry out the fine adjustment for minimizing the DC off-sets .delta..sub.C1 and .delta..sub.S1, the phase shift error .theta. of the quadrature carrier, and the carrier leaks .delta..sub.C2 and .delta..sub.S2, or the selection of the well balanced parts. However, such a fine adjustment or the selection of the well balanced parts requires a highly skilled operation, a time for making the adjustment, and an expensive measurement devices.
In order to solve the above described problems of the conventional quadrature modulator, there has been a proposition to use a linear transformation of the baseband signal inputs for the sake of the compensation of the linear distortion due to the DC off-set, as described in "Direct Conversion Transceiver Design for Compact Low-Cost portable Mobile Radio Terminals" by A. Bateman and D. M. Haines, 1989 Vehicular Technology Conference Proceeding, pp. 57-62, San Francisco, May 1989.
However, in the quadrature modulator described in this reference uses IQ detectors for extracting the in-phase component and the quadrature component of the output of the quadrature modulator in order to derive the linear parameters to be used in the linear transformation, so that it is necessary in this quadrature modulator to provide the IQ detectors having the higher accuracy than the quadrature modulator, but this in turn requires the fine adjustment of the DC off-set, amplitude balance, and orthogonal phase relationship for the IQ detectors themselves.
Furthermore, these conventionally proposed quadrature modulator configurations have a common problem that, since they are designed to deal with a particular type of the linear distortion alone, when the linear distortions due to the DC off-set and the amplitude imbalance are compensated, the linear distortion due to the inaccurate orthogonal phase relationship cannot be compensated, or visa versa.